Phase-locking system for local oscillators of tracking receivers



1968 R. E. GRAVES ETAL 3,419,

PHASE-LOCKING SYSTEM FOR LOCAL OSCILLATORS OF TRACKING RECEIVERSOriginal Filed Nov. 13, 1962 Sheet of 5 r ECEIVER i A RECEWER bvNo.

Y Y A RECUVER TRANSMlTlER- SYNC. SYNC.

DATA DROLESSOR i RANGE EEARlNG REE RE REF.

Nc, CARPAER 5MB CARRHER 5055 5. JGAIQA v5 o/v cos I 5 JACOB M. SAC/(SINVE 0R5 AGENT Dec. 31, 1968 v V S ETAL 3,419,814

PHASE-LOCKING SYSTEM FOR LOCAL OSCILLATORS OF TRACKING RECEIVERSOriginal Filed Nov. 13. 1962 Sheet 2 or 5 AOOMC CARRIER 404 MCSLABCARRIER I "I *fil 32 l I I RF I I F I AMPLIFIER I I AMPLIFIER I SUE:I I CARRIER 29a, I I CARRIE; I LOOP 4M6 I LOO 50' 55 OUTPUT I I 35 IENVELOPE vco I I vco ENVELOPE I DETECTOR 4M0 I I 4OOMC DETECTOR I I 53 7I I I L 2 2.7 I FILTER I I FILTER AMP. I I a AMP I I I L I 'L 1 REE---'-"-I I I I I 2 PHASE. I 42 I REFERENCE I COMPARATOR-4 I IOSCILLATOR: a, FILTER I I m; I 45 I L J I 35 I- I "2;; (400 2. 3| MC) I36 R F P I I I I I PHASE. I OR SYNC I I AMPLIHER I COMPARATOR I 397.69MC. I I EPFILTER I 555 EWNQ I I LOOD I I 2.31 Mc 5 R I /37 V I I vcoENVELOPE I 2.5mm DETECTOR I I 22 I FILTER I I I EI- AMD I I I I REF. I lL I R055 E. GRAVES DON M. JACOB JACOB M. SAC/(5 2 INVENTORS AGENT UnitedStates Patent 3,419,814 PHASE-LOCKING SYSTEM FOR LOCAL OSCIL- LATORS OFTRACKING RECEIVERS Ross Edwin Graves, Pacific Palisades, Jacob MiltonSacks, Palos Verdes Estates, and Don Murdock Jacob, Los Angeles, Calif.,assignors to TRW Inc., a corporation of Ohio Original application Nov.13, 1962, Ser. No. 237,229, now Patent No. 3,308,380, dated Mar. 7,1967. Divided and this application July 22, 1966, Ser. No. 567,269

6 Claims. (Cl. 331-2) ABSTRACT OF THE DISCLOSURE This inventiondiscloses a technique for phase tracking on incoming high frequencysignal by generating a local oscillator signal differing from theincoming signal by given low audio frequency, for example, 30 kc. Thelocal oscillator signal is injected ahead of any reactive elements, suchas an RF stage, thereby causing both incoming and injected referencesignal to be simultaneously affected and varied by reactive elements.Phase-locking of the incoming signal is achieved by a phase-locked loopon the 30 kc. offset audio frequency, which is not substantiallyaffected by the same previously mentioned reactive elements.

This application is a division of c-opending application Ser. No.237,229, filed Nov. 13, 1962 and now Patent No. 3,308,380.

This invention relates generally to a phase-stable receiver for use incontinuous wave (CW) Doppler systems.

One of the major problems in the development of precision CW guidanceand tracking systems is the design and implementation of the trackingreceivers. These receivers typically are required to operate over adynamic range of 80 db or more while tracking a vehicle traveling at avelocity up to 40,000 ft./sec. with an acceleration as high as 1,000ft./sec. The receivers must generally operate down to signal levels assmall as 140 dbm while tracking a vehicle with the aforementionedcharacteristics. With guidance and tracking systems of the sortcontemplated, position and rate measurements are made by means ofinterferometric techniques, employing system baselines whose lengths mayrange from a thousand feet to a number of miles. Rate measurements aremade by direct use of carrier Doppler data, while position measurementsrequire the use of subcarriers for the resolution of ambiguities -'inthe carrier phase differences measured by pairs of ground receivingstations (angular data) and in the round-trip phase shift fromground-to-vehicle-toground (range data). One of the preferred methods isto insert the high frequency subcarriers as single sideband subcarrierson the system carrier while the ranging subcarriers, and possibly thelow frequency angle ambiguity resolving subcarriers, may conveniently befrequency (or phase) modulated onto the carrier. In some cases, only onesingle sideband subcarrier may be employed, and this subcarrier may beswept in frequency or stepped discretely frequency for the purpose ofboth range and angle ambiguity resolution.

Regardless of the particular manner in which the subcarriers aremodulated onto the carrier, the accuracy with which carrier andsubcarrier phase data may be obtained from the tracking receivers is ofparamount importance for the operation of the guidance or trackingsystem. In addition, synchronization of such long baseline systemsrequires the use of radio links along the baseline paths; and thereception of such synchronizing data requires additional phase-stablereceivers. The problem of phaseice stabilizing an ultra high frequency(UHF) or microwave receiver is rendered extremely diflicult by thecombination of requirements, which include large dynamic range, abilityto function down to a very low signal level, capability of tracking overa wide range of vehicle velocities and for very large vehicleaccelerations, high output signal-to-noise ratios, and very high phasestability, all of which must be satisfied simultaneously. It should beobserved that phase accuracy requirements on such receivers may be asstringent as 0.1 electrical degree, while accuracies of a few electricaldegrees are required even in those systems for which the phase dataaccuracy requirements are more lenient.

This invention is concerned with techniques for receiver phasestabilization. The receiver design utilizes techniques which areapplicable to any widebase guidance interferometer or to any receivingsystem having comparable requirements. The receiving system iscompatible with both UHF and microwave frequencies; in the microwavecase the ferquency of the single sideband subcarrier might be as high asseveral hundred megacycles per second.

In this invention the received signals, which may include a carrier, asingle sideband subcarrier, and a synchronizing signal, are each trackedindividually in a separate receiving channel. The signals from the threereceiving channels are combined to yield the output phase data required.The receiver, actually constructed and incorporating the benefits ofthis invention, is capable of tracking the carrier alone down to 140 dbmand both the subcarrier and the carrier from -40 dbm to dbm, with amaximum source velocity of 140,000 ft./ sec. and a maximum sourceacceleration of 30 gs. The required tracking accuracy was 5 degrees forthe carrier and 0.1 degree for the subcarriers. The phase shifts of theoutput signals are kept small by using the principles described in thisinvention.

The receiver does not employ a conventional RF mixer and intermediatefrequency (IF) amplifier with automatic gain control (AGC). Instead,phase shifts in the high gain amplifier stages are minimized byinjecting a voltage controlled oscillator (VCO) signal, or what wouldnormally be the local oscillator signal. This injected signal is fed tothe input of the RF amplifier at a level that is large relative to thereceived signal and with a prescribed audio frequency offset from thereceived signal which is small relative to the bandwidths of theamplifying stages. The term audio frequency as employed here and in thefollowing discussion is used to emphasize that the offset frequency islow relative to the intermediate frequency which would be employed in aconventional receiver for this portion of the frequency spectrum. Invarious embodiments this offset frequency could range from as low asseveral kilocycles to as high as several megacycles. The sum of thesesignals, after amplification in the RF amplifier, is detected to obtainan audio frequency beat note (30 kc. in the preferred embodiment). Theaudio beat note (30 kc.) contains the desired phase informationoriginally on the incoming signal, which is further amplified prior tophase detection with respect to the output of a reference audiooscillator (30 kc.) to obtain an error signal to control the VCO. Forcertain applications it will prove desirable to heterodyne the output ofthe RF amplifier to an intermediate frequency for further amplificationprior to detection, or even to delete the RF amplifier entirely andperform injection of the VCO signal immediately ahead of the mixeremployed to heterodyne the composite signal to the aforementionedintermediate frequency. Use of an IF amplifier in the manner describedretains the basic advantages of the RF reference injection technique. Bythis technique, any phase shifts which are introduced in the receivedsignal by the RF amplifier are compensated by virtually identical phaseshifts on the injected RF reference signal, which is offset by theprescribed audio frequency (30 kc.). The result is that any phase errorintroduced on the received signal in the RF and in the IF amplifier, ifemployed, is substantially removed by beating the signal against theinjected reference in the detector. In the preferred embodiment it waspossible to increase the signal to a level 6 db above the injectedreference before significant phase shift occurred, which permittedattainment of additional dynamic range in the receiver for strongsignals.

In addition, the injected reference signal power is large relative tothe total noise power in the bandwidth presented to the detector, withthe result that the signal-tonoise ratio is not degraded bynoise-cross-noise products generated in the detector.

Further objects and advantages of this invention will be made moreapparent by referring now to the accompanying drawings wherein:

FIG. 1 is a block diagram of a long base line CW Doppler system forlocating and tracking objects in space;

FIG. 2 is a block diagram of a receiver illustrated in the system ofFIG. 1;

FIG. 3 illustrates the frequency spectrum of the received signals andthe injected signals used in the receiver of FIG. 4; and

FIG. 4 is a more complete block diagram of the system illustrated inFIG. 2.

The invention is more properly concerned with a phase stable receiver;however, in order to more fully appreciate the application and use ofthe receiver the invention is also described in connection with a CWsystem. requiring a phase stable receiver.

Referring now to FIG. 1, there is shown a continuous wave Doppler systemcomprising a transmitter 10 and a plurality of remote receivers 11, 12,and 13. The continuous wave Doppler transmitter 10 is arranged totransmit a CW signal to a moving object 14. The moving object may be asatellite or missile and may include a coherent transponder, which isone that receives the transmitted signal from transmitter 10 andrebroadcasts this signal with a frequency offset coherent with thereceived signal. Alternatively, the moving object may be passive innature so that it will be simply reflect any transmitted signal thatimpinges upon its surface. The reflected or doherent transmitted signalfrom the moving object 14 is adapted to be received by all of thereceivers 11, 12, and

13. The transmitter 10 is also adapted to transmit a synchronizing(SYNC) signal to all the receivers 11, 12, and 13, which SYNC signal isphase coherent with the CW signal transmitted to the moving object 14.The output signals from each of the receivers 11, 12, and 13, tOgethelwith frequency and timing (phase) information from the transmitter 10,are fed to a data processor 15, consisting of analog and digital dataextraction equipment and of computers and associated equipment, which isresponsive to the output of said receivers for determining range andbearing information of the moving object 14. Each of the receivers 11,12, and 13 is identical and is arranged to compare the phase of thecarrier signal received from the moving object 14 with the SYNC signalreceived directly from the transmitter 10. The phase change in each ofthe receivers is an indication of the relative movement object 14. Thethree receivers in this configuration are known as an interferometer andby themselves will produce sufficient information to determine pairs ofrange differences. In the case where the distance to the moving object14 is large relative to the separations (baselines) between thereceivers 11, 12, and 13, these range differences are substantiallyequivalent to angular position (bearing) information. The combination ofthe transmitter 10 with any of the receivers will produce ranginginformation which, together with the range difference information fromthe three receivers connected as an lnterferometer, will produce rangeand bearing information sufficient to track an object in space.

In one embodiment of the transmitter 10 transmlts a CW carrier signal of400 mc. with a subcarrier of 404 mc. Each of the receivers 11, 12, and13 receives reflections of the 400 mc. carrier and 404 me. subcarrierfrom the moving object 14. In this case, the phase data of interest arethe phase of the 400 mc. carrier and the phase of the 4 me. subcarrierwhich is obtained by demodulat ing the 404 mc. sideband with respect tothe 400 mc. carrier. In the following discussion, it will be convenientto refer to both the 4 mc. and the 404 mc. signals as subcarriers, withthe understanding that the 404 mc. signal is to be thought of asobtained by single-sideband modulation of the 4 mc. signal onto the 400mc. carrier. The phases of the 400 mc. received carrier signals and thephases of the received 4 mc. subcarrier signals are compared in pairs inthe data processor 15 to determine the range differences previouslydiscussed. Similarly, comparison of the phase of the 4 mc. subcarrierreceived from the moving object 14 by receiver 12 with the phase of the4 mc. subcarrier transmitted from transmitter 10, effected in the dataprocessor 15, permits the measurement of (ambiguous) fine range to themoving object 14. The SYNC signal generated by transmitter 10 was fixedat 2.31 mc. below the carrier signal of 400 me. Each of the receivers isarranged to track the 400 me. carrier signal, the 4 mc. modulating(subcarrier signal, and the 2.31 mc. SYNC signal. In a second embodimentof the system described in FIG. 1, the moving object 14 contained a 400mc. transmitter and 4 me. modulating source. The transmitter 10 wasarranged to transmit a 17/16 X400 mc. CW signal to the moving object 14,which then received the 17/16 X mc. carrier signal and phase coherentlyretransmitted the 400 mc. carrier that was modulated by the 4 mc. sourcelocated in the moving object. In this embodiment the ground basedreceivers 11, 12, and 13 performed the same operations as described forthe first embodiment. In this case, the received signals from the movingobject 14 were at a substantially higher signal level, therebymaterially improving the signal-to-noise ratio at the receiver. Thissystem is considered more desirable for the guidance and tracking offriendly objects where suitable transponding equipment may be installed.The principles of the phase stable receiver to be described are the samein either case. It should be observed, however, that when the 4 me.signal is obtained from a source in the moving object 14, the resultantreceived 4 mc. phase data cannot be employed for range measurement, as aconsequence of the lack of a 4 mc. phase reference in the ground system,even though they are completely satisfactory for measurement of rangedifferences.

The phase-stable receiver receives a carrier signal and modulatingsignal from the moving object and measures the phase change in themodulating signal by using the carrier signal as a reference. This phaseinformation when received from at least three different ground stationsis sufficient to give range difference information. In the preferredembodiment a SYNC signal containing the phase information of thetransmitted carrier is also sent to all ground stations in order toobtain range and angle information. By using a carrier frequency of 400me. and a modulating frequency of 4 mc., it can be shown that every 360degree phase change of the 400 mc. carrier is equivalent to 1.23 feetand that every 360 degree phase change of the 4 mc. modulating signal isequal to 123 feet.

It is possible by conventional techniques to measure the phase of the400 mc. carrier signal to within 5 degrees and the phase of the 4 mc.subcarrier to approximately /2 degree. The accuracies claimed for thesystem will therefore approximate 0.018 feet for the carrier signal and0.17 feet for the subcarrier signal. In other words, the 4 mc.subcarrier signal provides unambiguous range information for every 123feet measured by the 400 mc. carrier signal. Additional range ambiguitycan be resolved by amplitude modulating the carrier with a suflicientlylow frequency depending upon the maximum range expected. For example,using a 137 cycle signal will provide an unambiguous range of 40,000feet.

Referring now to FIG. 2, there is shown a phase-stable receiver adaptedto receive a 400 mc. carrier and a 404 mc. subcarrier signal. Thereceiver comprises a carrier loop 20, a subcarrier loop 21, and asynchronizing loop 22. The improved phase characteristics claimed forthe phase-stable receiver are believed due primarily to the use of an RFinjected reference signal whose frequency is offset by an audiofrequency from the frequency of the received signal. The offset audiofrequency was selected in the preferred embodiment to be 30 kc. Theinjected reference signal is amplified together with the carrier signaland detected to obtain the resultant audio frequency between the carriersignal and the injected reference signal. This technique reduces theeffect of phase shifts within the receiver to a very small value sincethe phase information is contained in the 30 kc. audio offset signal. Inthe carrier loop 20, the received 400 mc. carrier signal from antenna 23is fed to an adder 24 where the carrier signal and the injectedreference signal from VCO 25 are combined. Both the 400 mc. carrier andthe injected reference signal at a frequency of 400 mc.-i-kc. are fed toand amplified in an RF amplifier 26. The output of RF amplifier 26 isfed to an envelope detector 50'. The detected 30 kc. beat note is fed toa phase detector 27 where the phase of the 30 kc. audio offset iscompared with a reference signal generated by a free running 30 kc.oscillator 28. The output of the phase detector 27 is a DC signal whichis fed to a filter and amplifier 29 and, then, used to control thefrequency of the 400 me. VCO 25. The loop circuit just describedgenerates an output signal that is offset from the incoming carriersignal by 30 kc. In other words, the carrier loop circuit 20continuously tracks the difference or 30 kc. audio offset signal,thereby insuring an injected signal that has a fixed frequency offsetfrom the incoming carrier signal.

The subcarrier loop 21 comprises a mixer stage 29a. which is arranged toreceive the output signal from the VCO 25 and an output signal from a 4mo. VCO 30. The output from mixer 29 represents the injected referencesignal for the subcarrier loop 21 and is fed to an adder 31, where theinjected reference is combined with the received 404 mc. subcarrier fromantenna 23. The output from adder 31 is amplified by an RF amplifier 32and consists at least of the incoming subcarrier at 404 mc. and theinjected reference signal at 400.030-1-4 mc. or 404.030 mc. It can beseen therefore that in the subcarrier loop 21 RF amplifier 32 isactually amplifying two signals that are only 30 kc. apart, therebyminimizing the phase shift in the loop to a very small value. The outputof the RF amplifier 32 is fed to an envelope detector 53. The detected30 kc. beat note is fed to a phase detector 33 which compares the phaseof the detected 30 kc. beat note with the same reference signalgenerated by the 30 kc. oscillator 28. The output of phase detector 33is fed to a filter and amplifier 34, the output of which is used tocontrol the frequency of the 4 mc. VCO 30. The output of the subcarrierloop 21 is taken from the output of the 4 mc. VCO 30, which contains thenecessary phase information.

For those systems requiring only relative angle information, it is onlynecessary to compare the phase of this 4 mc. signal against thatreceived by other receiving stations to thereby obtain range differencedata for the moving object. For those systems requiring specificlocation of the moving object, it is necessary to compare the phase ofthe 4 mo. output signal with the phase of the transmitted subcarriersignal to thereby obtain range information. The combining of the rangeinformation together with range difference data will supply thenecessary information to locate the moving object in space.

The SYNC carrier loop 22 supplies the phase information which is relatedto the transmitted 400 mc. carrier signal for providing the system withthe means for obtaining range and angle information. The 400-2.31 mc.SYNC signal is received from the transmitter by means of antenna 35 andfed to adder 36. The injected reference signal is composed of a firstsignal from the output of the 400 mc. VCO 25, which is mixed with theoutput of a 2.31 mc. VCO 37 in mixer 38. The signal from the VCO 25 isactually 400.030 mc., which is mixed with the 2.31 mc. output from VCO37 in mixer 38. The injected frequency will therefore be 397.72 mc.,which will be added to the received 397.69 mc. SYNC signal, whichsignals are amplified by RF amplifier 39. The output of the RF amplifier39 is fed to an envelope detector 56. The detected 30 kc. beat note isfed to a phase detector 40 where the 30 kc. beat note frequency is phasecompared with a reference signal received from the same 30 kc.oscillator 28. The output of the phase detector 40 is fed to a filterand amplifier 41, the output of which controls the frequency of the 2.31mc. VCO 37. The 2.31 mc. output frequency contains phase information ofthe 400 mc. carrier and when used in conjunction with the 4 mc. VCO 30will produce range and bearing information. A review of the carrier loop20, the subcarrier loop 21, and the SYNC loop 22 will show that allthree loop circuits are locked together and actually lock on the 30 kc.audio offset frequency.

Since each of the defined loop circuits actually tracks a 30 kc. signal,it is important that the phase shift of each loop circuit be the same orvery close to being the same as dictated by the ultimate requirements ofthe overall system. By using appropriate loop gains in the carrier loop20, subcarrier loop 21, and the SYNC loop 22, the dynamic phase errorresulting from acceleration of the moving object may be made nearlyidentical in all of the defined loop circuits. By keeping the phaseerror the same for all loops, no resultant error will appear in theoutput signal. Phase differences between the loops, caused bydifferential phase shifts in the different tracking loop filtercircuits, phase detectors, and filter amplifiers, can be reduced byphase detecting circuits 42 and 43. Phase detecting circuit 42 comparesthe phase of the 30 kc. beat note signal generated in the subcarrierloop 21 with the phase of the 30 kc. beat note signal generated in thecarrier loop 20 by feeding the signals directly to a differencer 44. Theoutput of the differencer 44 is fed to a phase comparator and filter 45,where the signal is compared with a reference 30 kc. signal generated bythe 30 kc. oscillator 28. The output of the phase comparator 45 is fed tthe filter and amplifier 34 in the subcarrier loop 21, thereby insuringthat any phase difference of the 30 kc. beat note signal generated inthe carrier loop 20 will be the same as that of the 30 kc. beat notesignal generated in subcarrier loop 21. In a similar fashion, the phaseof the 30 kc. beat note signal generated in the SYNC loop 22 is comparedwith the phase of the 30 kc. beat note signal generated in the carrierloop 20 by means of phase detecting circuits 43. The 30 kc. beat notesignal from carrier loop 20 and the 30 kc. beat note signal generated inthe SYNC loop 22 are fed to a differencer 46. The output of thedifferencer 46 is fed to a phase comparator and filter 47, Where thesignal is compared against a reference signal generated by the 30 kc.oscillator 28. The output of the phase comparator 47 is fed to thefilter and amplifier 41 in the SYNC loop 22, thereby insuring that thephase of the SYNC loop and the carrier loop will be the same.

The phase detecting circuit-s 42 and 43 are preferably commutated phasedetectors having high gain, low error and low drift. Circuits of thiscaliber have been disclosed and claimed in copending application,entitled Precision Phase Detector, Ser. No. 226,118, now Patent No.3,142,- 804. The feedback signal from phase detecting circuit 42 couldhave been alternatively directed to the filter and amplifier 29 in thecarrier loop 20. This feedback results in a reduction of the initialphase error obtained without the phase detecting circuits by a factorwhich depends on the grain of the phase detecting circuit.

Referring now to FIG. 3, there is shown a frequency spectrum unaffectedby Doppler shift, illustrating the received carrier signal at 400 mc.and the subcarrier displaced from the carrier by 4 mc. Also shown is thereceived SYNC signal which is displaced from the carrier on the low sideby 2.31 mc. As mentioned previously, the injected reference signals forthe carrier signal, the subcarrier signal and the SYNC signal are eachshown displaced from their respective received signal by the audiooffset frequency of 30 kc. The relative amplitudes of the injectedreference signals are shown in an exaggerated condition to more fullyillustrate the increased amplitude of the injected reference signalsover the corresponding received signals.

Referring now to FIG. 4, the-re is shown a block diagram illustrating inmore detail the system described in FIG. 2. In describing the moredetailed block diagram of FIG. 4, similar numbers used in connectionwith FIG. 2 will be used whenever the complete block performs the samefunction as described in FIG. 2. However, wherever additional blockshave been added to more fully describe the function of the system, newnumbers will be assigned and used.

In both FIGS. 2 and 4, the 400 mc. and 404 mc. signals are illustratedas being received and separated prior to addition of the injectedreference signals for purposes of illustration only and to helpunderstand the invention. It should be noted that whenever two receivedsignals are so close together that the indicated power splitter willproduce a 3 db loss in the desired signal component, there- 'bydegrading the noise performance of the receiving system, this loss maybe eliminated by inserting the injected reference signals before anypower splitting and then amplifying as shown before envelope detectingthe signals.

The incoming carrier signal fed to adder 24 is actually a frequencyvarying signal having phase information which may be represented as:

The injected reference signal generated by the VCO 25 is locked to theincoming carrier signal and offset therefrom by 30 kc. as previouslydescribed and may be mathematically represented as:

Both the incoming carrier signal and the injected reference signal areamplified by the RF amplifier 26. The difference signal containing thephase information is detected by an envelope detector 50. The output ofthe envelope detector 50 is the 30 kc. offset signal which is fed to afilter 51 to remove the high frequency carrier components. The filtered30 kc. signal from filter 51 is fed to a limiter 52 for generating a 30kc. signal at a given amplitude. In the embodiment described the definedlimiters 52, 55, and 58 must be extremely phase stable with respect toamplitude. Details of a limiter circuit having the necessary phasestability are described and claimed in a copending application, Ser. No.237,267, filed Nov. 13, 1962, now abandoned, and assigned to the samecommon assignee. It should be understood, however, that while thelimiter technique for performing automatic gain control on the 30 kc.signals represents the preferred embodiment, there are many othertechniques that are also suitable. For example, the simple expedient ofusing an AGC audio amplifier having a phase shift that varies a smallamount with changes in AGC bias. Phase detector 27 compares the phase ofthe reference 30 kc. oscillator 28 with the limited 30 kc. signal fromlimiter 52 and generates a pulsating DC signal .of the proper amplitudeand sense as a function of the phase difference between the input 30 kc.signals. The pulsating DC output from the phase detector 27 is fed tothe filter and amplifier 29 which generates a DC signal which is used tocontrol the output of the 400 mc. VCO 25. The output of the VCO 25 willbe the injected reference signal having a frequency of 400.030 mc. and aphase given by:

A review of the subcarrier signal tracking loop 21 will show that the404 mc. subcarrier identified as w t+ is fed to the adder 31 togetherwith the injected reference signal which is 30 kc. offset from thesubcarrier and identified as (w t+w t)|( In a similar fashion, asdescribed for the carrier loop 20, both signals are amplified by an RFamplifier 32 and fed to an envelope detector 53. The detected 30 kc.signal from the envelope detector 53 is fed to a filter 54, whichremoves the high frequency carrier components and then is fed to alimiter 55. The output of the limiter 55 is compared with the output ofthe 30 kc. oscillator 28 in the phase detector 33. The pulsating outputfrom the phase detector 33 is fed to a filter and amplifier 34, whichgenerates a DC signal varying in amplitude and sense as a function ofthe difference in phase between the reference signal and the output oflimiter 55. This DC signal controls the 4 mc. VCO 30. The 4 me. outputfrom VCO 30 is fed to the mixer 29, which combines the 4 me. signal withthe 400.030 mc. output from the VCO 25, located in the carrier loop 20.The injected reference signal fed to the adder 31 is therefore a 404.030mc. signal, which is 30 kc. offset from the received 404 me. subcarriersignal, which is also fed to the adder 31.

The SYNC loop 22 is very similar to the subcarrier loop 21 in that theSYNC signal is received as a subcarrier that is 2.31 mc. removed on thelow side of the 400 mc. carrier. This received signal is fed to theadder 36, which also receives the injected reference signal from themixer 38. Both signals are amplified by the RF amplifier 39 and fed toan envelope detector 56 where the difference signal of 30 kc. isextracted. This signal is fed to a filter 57 and a limiter 58 in thesame fashion as previously described. The phase detector 40 generates apulsating DC signal in response to the phase difference between the 30kc. signal received from the limiter 58 and 30 kc. reference oscillator28. The pulsating DC signal is fed to the filter and amplifier 41, theoutput of which controls the 2.31 mc. VCO 37. This 2.31 mc. signal iscombined with the 400.030 mc. signal from the VCO 25 and the carriertracking loop 20 in the mixer 38. The resulting injected referencesignal of 397.72 mc. is combined in the adder 36 with the received SYNCsignal of 397.69 mc.

The advantage of the defined receiver is that any phase shiftsintroduced into the carrier, subcarrier, and SYNC signal receivingchannels due to Doppler shifts or signal level variations in the RFamplifier and limiter amplifiers, which are of the same magnitude forboth channels, will not appear as phase errors in either the 4 mc.subcarrier output signal or in the 2.31 mc. carrier output signal. Afurther advantage is that the dynamic phase error due to acceleration ofthe moving object will tend to cancel between the carrier and subcarrierchannels (or carrier and synchronizing channels). The dynamic phaseerror which can result from acceleration of the moving object can beminimized by the proper selection of reference gains in the carrier loopcircuit and the subcarrier loop circuit. By making the dynamic phaseerror exactly the same for each loop, no resultant error will appear inthe output signal. In an idealized situation the phase error of eachloop can be made identical; however, it is known that active elements inthe loop filters, phase detectors and loop filter amplifiers will causedifferential phase shifts. It is the purpose of phase detecting circuit42 to compare the phase shift in the 30 kc. signal in the carrier loop20 with the phase shift of the 30 kc. signal in the subcarrier loop 21and in response thereto generate an additional 9 DC signal of properamplitude and sense to thereby make the phase shift of the subcarrierloops 21 the same as the phase shift in the carrier loop 20. The phasedetecting circuit 43 performs the same function with respect to thecarrier loop 20 and the SYNC loop 22, thereby insuring that all threeloop circuits will have substantially the same dynamic lag phase error.In this manner dynamic lag errors in the outputs of the 4 mc. VCO 30 andthe 2.31 mc. VCO 37 are substantially eliminated. The phase comparisoncircuit 42 comprises a difference 44 which receives a first inputconsisting of the limited 30 kc. signal from the output of limiter 52,located in the carrier loop 20, and the limited 30 kc. signal from thelimiter 55, located in the subcarrier loop 21. The two 30 kc. signalswhich are of approximately equal magnitude are actually added inantiphase by the differencer 44 prior to being filtered in a bandpassfilter 59. The output of the filter 59 is then amplified by an amplifier60 and phase detected with respect to the output of the 30 kc.oscillator 28 in a phase detector 61. Since the phase detector is onlysensitive to a voltage that is in phase quadrature with respect to theinitial input signals, it can be shown that the quadrature voltage willarise only when a phase difference exists between the two input signalsand will not exist when there is only an amplitude difference. In otherwords, should be angle between the original input voltage and the phasedetector reference voltage not to be 90 degrees, then any difference inthe amplitudes of the signal component from limiter 55 in the subcarrierloop 21 and that from limiter 52 in the carrier loop 20 will produceonly a second order effect on the output error signal from phasedetector 61. In the preferred embodiemnt, the defined second ordereffect is reduced or eliminated by controlling the output amplitude ofthe last stage of the limiter 55 with an AGC circuit which operates tomaintain the amplitude of the signal component of the output of thislimiter substantially equal to the amplitude of the signal component ofthe output of limiter 52. Alternately, this defined second order effectcan be reduced or eliminated by means of external circuitry for servoingthe phase error so that it is always 90 degrees with respect to thephase detector reference voltage. The pulsating DC signal from the phasedetector 61 is fed to a filter 62 which generates a DC signal varying inamplitude and sense as a function of the phase difference between thecarrier loop 20 and the subcarrier loop 21. This DC signal is fed to thefilter and amplifier 34 where it is combined with the DC control signalgenerated by the subcarrier loop 21, which is ultimately used to controlthe 4 me. VCO 30. In this manner the dynamic phase error of thesubcarrier loop will be the same as the phase error of the carrier loop.The phase comparing circuit 43 is similar in function to the phasedetecting circuit 42. The phase of the carrier loop 20 is compared withthe phase of the SYNC loop 22 in order to generate an error signal tothereby cause the SYNC loop to have the same phase lag as the carrierloop 20. The 30 kc. signal from the output of limiter 52 in the carrierloop 20 and the 30 kc. signal from the output of limiter 58 in the SYNCloop 22 are both fed to the differencer 46 located in the phasecomparing circuit 43. The difference betwen these two signals representsan error signal indicating a phase difference between the output of thecarrier loop and that of the SYNC loop. This error signal is fed to afilter 63, which in turn feeds an amplifier 64. The output of amplifier64 is compared with the output of the 30 kc. reference oscillator 28 bymeans of phase detector 65. The pulsating DC output from the phasedetector 65 is fed to a filter 66 which generates a DC signal varying inamplitude and sense according to the phase difference between the outputof the carrier loop and that of the SYNC loop. The output of the filter66 is fed back into the filter and amplifier 41, located in the SYNCloop 22. This feedback DC signal is combined with the DC signalcontrolling the 2.31 mc. VCO 37 to thereby correct the phase of the SYNC10 loop output from the 2.31 mc. VCO 37 for dynamic lag phase errors inthe carrier loop.

Considerable reduction in the effect of drift in the output of the phasecomparing circuits 42 and 43 is obtained as a result of the additionalgain from amplifiers and 64, which can be inserted in the signal pathprior to the phase detectors without causing saturation. Insertion ofthis additional gain is possible because of the reduction in inputamplitude which is obtained by adding the signals from the limiters 52and 55 in antiphase and by adding the signals from the limiters 52 and58 in antiphase. In addition, low drift and very small noise unbalanceof the phase detectors can be obtained by means of the commutationtechnique described in said copending application. The 30 kc. referenceof the commutated phase detector may be inverted in phase every 16cycles, thereby producing a 940 c.p.s. error signal at the output of thephase detector. The peak-to-peak amplitude of this error signal isproportional to the phase error between the two initial input signalsfrom the limiters, and the AC component of the phase detector output hasone of two phases, depending on the sign of the phase error between thesignals from the two limiters. The AC component of the phase detectoroutput is amplified and then decommutated in a synchronous clampingcircuit. The complete circuit, including the 30 kc. referencecommutator, the simple phase detector, and the decommutator is referredto as the commutated phase detector. This commutated phase detector is avery sensitive device having extremely low drift. Because of the highgain employed, the output signal will saturate for only a few degrees ofphase difference in the input signals. However, only a few degrees phaseerror will ever occur at the input to the commutated phase detectorsduring normal operation of the signal combiner loops, as embodied in thephase comparing circuits 42 and 43.

The practice of adding the two signals at the outputs of the limiters inantiphase and phase detecting the resultant sum with respect to theoutput of the audio reference oscillator, rather than merelyphase-detecting the output of one of the limiters with respect to theoutput of the other limiter, is employed in order to reduce the level atwhich thresholding occurs in the phase detector 61 or 65. If the outputof one limiter 52 were merely to be phase detected with respect to theoutput of the other limiter 55, the action of the phase detector wouldgenerate noise-cross-noise products, caused by the noise in one channelbeating with that in the other channel, which would cause the phasedetector to threshold for input signal-to-noise ratios of the order ofunity or somewhat below. Moreover, in this case it would not be feasibleto perform substantial additional filtering in either or both of the twochannels prior to phase detection because of the phase shifts and driftswhich would be caused by the required audio filters.

In the present invention the performance of the phase detector isimproved since the noisy signal obtained by adding the outputs of thepair of limiters in antiphase is phase detected with respect to anoise-free signal from the reference audio oscillator. This fact impliesthat, as long as the level of the reference signal is large relative tothe total level of the signal-plus-noise in the other phase detectorinput channel, thresholding caused by noisecross-noise products will notoccur. In addition, after the two noisy signals have been added inantiphase, it is feasible to employ additional gain before phasedetection and also to use a narrowband filter to improve thesignalto-noise ratio presented to the phase detector in the signalchannel. Any phase shift introduced by the narrowband filter on thesignals from the two limiter-amplifiers will be identical and, hence,will produce negligible resultant offset in the phase detector as longas the amplitudes of the signal components in the limiter outputs arewell balanced.

This completes the description of the embodiment of the inventionillustrated herein. However, many modifications and advantages thereofwill be apparent to persons skilled in the art without departing fromthe spirit and scope of this invention. Accordingly, it is desired thatthis invention not be limited to the particular details of theembodiment disclosed herein, except as defined by the appended claims.

The embodiments of the invention in which an exclusive property orprivilege is claimed are defined as follows: 1. In a system having atleast two phase-locked loop circuits in which each loop circuit isadapted to phase lock on the difference between a received signal and aninjected signal differing in frequency by an arbitrarily selected offsetfrequency comprising:

means in each of said loops for detecting said offset frequency, meansfor adding said detected signals in phase opposition to produce an errorsignal, and means for controlling the phase of one of said signals withsaid error signal to minimize said error signal. 2. In a system havingat least two phase-locked loop circuits in which each loop circuit isadapted to phase lock on the difference between a received signal and aninjected signal differing in frequency by an arbitrarily selected offsetreference frequency comprising:

means in each of said loops for detecting said reference signal, meansfor adding in phase opposition said detected signals to produce an errorsignal, means for amplifying said error signal, and means for changingthe phase of said detected signal in one of said loops with a componentof said amplified error signal to minimize said error signal. 3. A phasecomparing system comprising: oscillator means for generating asubstantially constant frequency reference signal, at least two loopcircuits each comprising means for receiving a frequency varying carriersignal, means responsive to said reference signal for generating aninjection signal at a frequency differing from said frequency varyingsignal by said reference signal frequency, means for adding said carriersignal and said injection signal, means for detecting said referencesignal from said added signals, means responsive to the phase differencebetween said detected reference signal and said reference signalreceived directly from oscillator means for controlling said injectionsignal frequency, means for adding said detected reference signals fromsaid loop circuits in phase opposition with each other to produce anerror signal, and means for controlling one of said loop circuits withsaid error signal to minimize said error signal. 4. A phase comparingsystem comprising: an oscillaotr for generating a substantially constantfrequency reference signal, at least two loop circuits each comprisingmeans for receiving a frequency varying carrier signal, means responsiveto said reference signal for generating an injection signal at afrequency differing from said carrier signal by said reference signalfrequency,

means for combining said carrier signal and said injection signal,

means for detecting said reference signal from said combined signal,

means responsive to the phase difference between said detected referencesignal and said oscillator generated reference signal for controllingthe frequency of said injection signal,

means for adding said detected reference signals from said loop circuitsin phase opposition to produce a first error signal,

means for amplifying said first error signal,

means for comparing the phase of said amplified first error signal withthe phase of said oscillator generated reference signal to produce asecond error signal, and means for controlling one of said loops withsaid second error signal to minimize said first error signal.

5. A phase comparing system comprising:

an oscillator for generating a substantially constant frequencyreference signal,

at least a first, a second, and a third loop circuit, each comprisingmeans for receiving a frequency varying carrier signal,

means responsive to said reference signal for generating an injectionsignal at a frequency differing from said signal by said referencesignal frequency,

means for adding said carrier signal and said injection signal,

means for detecting said reference signal from said combined signal,means responsive to the phase difference between said detected referencesignal and said oscillator generated reference signal for controllingsaid injection signal,

means for adding said detected reference signals from said first andsecond loop circuits in phase opposition with each other to produce afirst error signal,

means for controlling said second loop circuit with said first errorsignal to minimize said first error signal,

means for adding said detected reference signals from said first andsaid third loop circuits in phase opposition with each other to producea second error signal, and

means for controlling said third loop circuit with said second errorsignal to mimimize said second error signal.

6. A system according to claim 5 in which said first and second loopcircuits are phase-backed loops adapted to receive a carrier andsubcarrier signal from a moving object, and said third loop is adaptedto receive a synchronizing signal from a ground source containing phaseinformation about said carrier signal.

References Cited UNITED STATES PATENTS 2,775,701 12/1956 Israel 331-2JOHN KOMINSKI, Acting Primary Examiner.

S. H. GRIMM, Assistant Examiner.

U.S. Cl. X.R.

